Ecler-PAM1000-pwr-sm维修电路原理图.pdf

PAM1400/1000/600/300 SERVICE MANUAL RadioFans.CN 收音机爱 好者资料库 SERVICE MANUAL PAM1400/1000/600/300 INDEX - BLOCK DIAGRAM - SCHEMATICS - COMPONENTS LOCATION SCHEMA - TESTING AND QUALITY CONTROL - TECHNICAL CHARACTERISTICS - WIRING DIAGRAM - MECHANICAL DIAGRAM - PACKING DIAGRAM RadioFans.CN 收音机爱 好者资料库 1 MODULE CIRCUIT 11.0504B OPERATION - DESCRIPTION The control element is the operational NE5534. This is a very low noise operational, especially designed for very high quality applications in professional audio equipment, control equipment and telephony channel amplifiers. The operational is internally compensated for a gain equal to or higher than three. Frequency response can be optimized with an external compensation capacity, for several applications (unity gain amplifier, capacitive load, slew-rate, low overshoot, etc.). Characteristics: Small-signal bandwidth: 10Mhz Output drive capability: 600 10V(rms) at Vs=18V Input noise voltage: 4nV/ % Hz DC voltage gain: 100000 AC voltage gain: 6000 at 10KHz Power bandwidth: 200KHz Slew-rate: 13V/s Supply voltage range: 3 to 20V POWER SUPPLY The BF871 and BF872 transistors are mounted in a common base configuration, in a current source structure. The current sources have a double function: polarizing the gate-source links in the MOSFETs to the limit of the conduction and moving the voltage variations at the operational output which are refered to ground to voltage variations refered to high voltage power supply. The polarization point is calculated so the voltage dropout in Rc (R112+R111) is the limit voltage of conduction of the MOSFETs (.2 to 3V), enough to carry the bias current. If we modulate in AC the base-emitter voltage, the Ic and VRc will vary proportionally. In our configuration, as the reference voltage Vref is constant (it is a part of the operational power supply), we add the operational output voltage to the transistors emitter through Re (R107-R108). The Rc value fixes the source output impedance. We do not recommend to raise it higher than 1K because of frequency response and slew rate reasons. This voltage circuits gain is, as usual in a common base configuration with Rc/Re emitter resistor, 0.45. 2 BIAS CURRENT ADJUST The bias current adjust is performed through the variable resistor connected between the emitters of the current sources R110 (5K). It delivers a supplementary current (it does not go through the operational) which simultaneously increases the voltage which falls in the Rc load resistors. This is the easiest way of acting with just one adjust over both branches at the same time. In order to adjust the bias current the adjustable resistor must be varied until a current of about 80mA circulates through each MOSFET. So, for instance, for a PAM1400 in which there are six MOSFETs it will be 80 x 6 = 480mA. The bias current depends on the MOSFETs temperature and the stabilizing circuit transistors temperature. TEMPERATURE STABILIZING CIRCUIT Temperature affects MOSFETs conduction in two different ways: first, the conduction threshold voltage has a negative temperature coefficient; second, the drain-source conduction resistance increases with temperature. Depending on which of the two things is predominating the temperature coefficient of the drain can be positive or negative. In our case, in which the gate-source voltage in the MOSFETs is very low when they conduct, the temperature coefficient of drain current -which is positive- is predominating. To avoid thermal runaway in the polarizing current we must decrease the gate-source voltage as the MOSFETs get hot. Temperature stabilization is performed by modifying the reference voltage of both sources. If the temperature increases the Vref must decrease so that Ic and VRc decrease and, as a consequence, the gate-source voltage also decreases. The circuit used is shown in figure 3. The base-emitter Vbe temperature/voltage feature is used to obtain the final result we need. The main idea is adequately choosing R1 and R2 to obtain the right temperature coefficient. 3 SYMMETRY ADJUST The threshold voltage varies much, even between MOSFETs of the same kind. When connecting them in parallel we must be careful that they all have the same conduction current if we want equal currents circulating in all of them. If the conduction voltage of P an N channels MOSFETs is not the same they will conduct different currents, even when we apply identical gate-source voltages. As the bias current of the N MOSFETs must be identical to the one of the P MOSFETs the feedback will correct the continuous voltage at the operational output to polarize the MOSFETs with different voltages until both conduct equal currents. If the operational output is not 0 V its capacity to give voltage and current is not the same in both senses. To avoid this we must put a symmetry adjust. It is just an adjust which allows to vary the collector resistance of one of the current sources (R111). The symmetry adjust does not correct the asymmetrical clipping saturation of the power amplifier with real load. This happens because the conduction resistors (Ron) of the MOSFETs N and P are not equal. Channel P has a higher Ron than channel N. This characteristic depends on the MOSFETs physical construction. POWER MOSFETs The MOSFETs used are IRFP9240 (P) and IRFP240 (N). They are assembled in a common source configuration so they can be completely saturated. This kind of configuration has two drawbacks compared to a common drain one: less stability (because of the configuration gain itself) and high output impedance in open loop. The source resistances (0.22) are needed for the MOSFETs to work in parallel. E.g.: Two MOSFETs excited by the same Vgs voltage (gate-source voltage) of 5V. If they have different transconductance curves (Id function Vgs) they will conduct different drain currents; lets say 1A and 3A. The second one will dissipate more power and will get hotter. The use of source resistances tends to match the current that each of the MOSFETs connected in parallel is conducting. 4 This resistance performs a negative feedback on the gate, lowering down the Vgs, relating to the drain current; like this: Vgs = Vgg - Id*Rs The higher the Id, the lower the Vgs voltage. The gate is protected by a zener, preventing a possible overload during an unexpected change from overload to real clipping. Given the high input impedance and the broad frequency response of the MOSFETs there is a high risk of self-oscillations if all gates are excited connected to the same node. Intercalating serial resistances and ferrite beads at the gate this possibility is minimized, because the Q of the LC network made by the inductances and gate-source capacity is reduced. PROTECTION CIRCUIT The protection circuit monitors the dissipated power at the MOSFETs stage. It has two basic parts: MOSFET Id current detection. MOSFET Vds voltage detection. The goal is limiting the MOSFET so it works inside an area close to the SOA, as indicated in the figure. We chose channel N because, due to construction reasons, its SOA is lower. ZONE A. This zone is for very low loads, around 0. As the load voltage is very low, the voltage held by the MOSFET will always be high. The protections should be activated with very low current. Fast protections and some of the slow ones are working in this zone. The circuit that configures the fast ones is made of: D120, D121, D123, R174, R175, R176, R177, R178, R179, C127, Q122 and Q123 for the N channel. There is also an equivalent circuit in the P channel. These start working when there is a sudden current variation because of a shortcircuit or a transitory. The reaction time -from the exact moment when these things occur to when the current stops circulating through the MOSFETs- is about 80s. The time constant is given by C127, R174 and R179 and the load circuit made by the LED diode of the IC104 (opto-coupler). Please note that in order for the protection to be activated Q122 and Q123 must conduct simultaneously, through which R174 is linked to negative power supply, being C127(1F) loaded very quickly through this resistance, activating the LED of the opto-coupler, sending a pulse to the protection circuit, which will open the corresponding channels relay, being this way the output from the power amplifier disconnected from the load (0), in this case. Q122, together with the zeners and the base polarization resistances, configure the voltage detector (this group is in parallel with the Vds voltage of the N MOSFET). Id CBA Vds 5 Q123, together with the resistances which make the base divider, configure the current detector (this divider takes its voltage from one of the source resistances of a N MOSFET, which is proportional to the current circulating through itself). The threshold separating zone A from zone B is determinated by the D125 zener. When this zener stops working and there is no current circulating through it because the Vds voltage is lower (lets remember this circuit is also in parallel with this voltage) or, what is the same, the load voltage grows because it is not 0O anymore and has a given value, like 0.5 to 1, and the help given by D126 stops so more current will be needed for the shot. We have climbed the first stair of the stairway of the SOA graphic. When the zeners D124 and D118 stop working because the load voltage goes on growing (values higher than 1) or -what is the same- the Vds decreases, the Q125 transistor does not receive current anymore in its base and so it is shorted, allowing Q124 to enter conduction. This way R172 stays in parallel with the base-emitter of Q121, making up a voltage divider with R173. This divider will climb another stair of the stairway and enter the ZONE C. The link between the modules protection circuit and the relays control circuit is made through IC103 and IC104 which are, as mentioned earlier, opto-couplers, just to insulate the existing high voltages at the power amplifying module, 90V in the case of the PAM1400, and the power supply voltage of the existing logic circuits in the relays control card. Once the pulse generated by the protections is detected, the control circuitry resident in the protection card, appart from opening the corresponding relay, returns the signal A.O. SUPPLY CONTROL to the module, which cuts by means of Q119, Q120 and IC102 the operationals power supply. This is the way to insure a fast and safe cut of the Id current in the MOSFETs (around 80s time), because they stop receiving their respective reference voltages and, consequently, their Vgs polarization voltages so they are cut. The circuit is shown in figure 9 and its operation is very simple. When the A.O. SUPPLY CONTROL (+10V) signal appears, the Q119 transistor starts conducting, shortcircuiting to ground the positive power supply of the operational. On the other hand, the signal is also applied to the IC102s LED (opto TIL112 (4N35), which puts its internal transistor and Q120 into conduction, connecting the negative power supply of the operational to ground. 6 7 8 ZOBEL NETWORK This circuit tries to get a constant load impedance for the power module, in spite of the amplifiers load and frequency, to avoid phase shifting of the feedback signal. The values have been experimentally calculated through a study with square signal by trying to minimize the power amplifiers ringing with very capacitive loads (2,2F/4). The Zobel Network eliminates possible oscillations of the MOSFETs between 5MHz and 10MHz, too. This is why it must be physically placed at the modules output, avoiding long wiring. Great care must be taken for the signal not to be too shifted at the output, because the feedback could turn negative. FEEDBACK The whole amplifier is compensated with just one capacity, which places the amplifiers general pole at: 1 Fp = - = 140KHz 2*Rf*Cf Rf = R106 Cf = C109-C110 9 PROTECTION CIRCUIT 11.0411 OPERATION - DESCRIPTION The circuit is configured by: - A POWER SUPPLY. - A THERMAL PROBE DC AMPLIFIER. - A TEMPERATURE DETECTOR. - A DC OUT DETECTOR PER CHANNEL. - A CLIP CIRCUIT PER CHANNEL. - A RESET (TURN OFF/TURN ON) CIRCUIT. - A BINARY COUNTER PER CHANNEL. - TWO MONOSTABLE CIRCUITS PER CHANNEL. The circuit power supply is performed through various sources: +V, modules power supply. This voltage feeds the relays circuit, manual reset circuit and part of the clip circuit. Alternate voltage from a transformers secondary (manual reset circuit). There is also a stabilized 10V power supply which feeds the cards circuitry, made of IC301 (7805) plus the zener D302 (Z4.7) 4.7+5 . 10V. We will also need a regulated power supply to get 14Vmax at 0.7A, which can be obtained with IC302 (7805) plus an auxiliary circuitry that will be analysed below. The cooling fan speed is automatically regulated in relation to the power modules temperature, which is read by a thermal probe (LM35D), jointly linked to the heat sink. This high sensitivity thermal probe gives variations of 10mV for every EC. This voltage is picked up and amplified by the IC305 (LM358). Of course, there is a probe for each L and R heatsink. The output of both amplifiers is linked through two diodes D304 and D305, making an O gate, whose cathodes go to the regulator, applying the DC of any of them to the regulator. This provides a variable voltage at its output which oscillates from a minimum of approximately 7V for a temperature of 20EC (cold heatsink) to a maximum of 14V for temperatures of 76EC or higher. The gain of the amplifiers has been calculated for this temperatures window. The maximum voltage allowed by the heatsink in order to work properly is 14V. This maximum is given by the zener D305 (Z9.1/1); as the regulator is a 7805 the voltage will be -as maximum- 9.1+5 = 14.1V. When the zener is not working (not enough voltage) the voltage on the fan will be the output amplifiers, less 0.6V (diodes fall), plus the 5V of the IC302. 10 11 TEMPERATURE DETECTOR This circuit is calculated to operate over the output relay opening it if any of both modules temperature excedes 90EC, approximately. It is made with a comparator per channel (L-R), resident in the same IC306. Both share a reference voltage provided by D306 (TL431A), which gives excellent stability at that voltage 1%. These comparators reveive, like the DC amplifiers, the signal from their probes, comparing them with the Vref. Once this voltage is surpassed by any of both probes, the output of the corresponding comparator is balanced to the power supply (+10V), acting through D307, R318, D308 and R319 over the respective bases of transistors Q301 y Q307, which makes the corresponding relay open. This output is also connected to the THERM